Transmission Line Distributed Oscillator

ABSTRACT

In one embodiment, an integrated circuit antenna array includes: a substrate, a plurality of antennas adjacent the substrate; and an RF network adjacent the substrate, the RF feed network coupling to a distributed plurality of amplifiers integrated with the substrate, wherein the RF feed network and the distributed plurality of amplifiers are configured to form a resonant network such that if a timing signal is injected into an input port of the RF network, the resonant network oscillates to provide a globally synchronized RF signal to each of the antennas.

RELATED APPLICATIONS

This application is a continuation-in-part of U.S. application Ser. No.11/182,344, filed Jul. 15, 2005, which in turn is a continuation-in-partof U.S. application Ser. No. 11/141,283, filed May 31, 2005. Inaddition, this application claims the benefit of U.S. ProvisionalApplication No. 60/721,204, filed Sep. 28, 2005.

TECHNICAL FIELD

The present invention relates generally to oscillators and moreparticularly to a transmission line distributed oscillator.

BACKGROUND

Conventional beam forming systems are often cumbersome to manufacture.In particular, conventional beam forming antenna arrays requirecomplicated feed structures and phase-shifters that are impractical tobe implemented in a semiconductor-based design due to its cost, powerconsumption and deficiency in electrical characteristics such asinsertion loss and quantization noise levels. In addition, such beamforming arrays make digital signal processing techniques cumbersome asthe operating frequency is increased. In addition, at the higher datarates enabled by high frequency operation, multipath fading andcross-interference becomes a serious issue. Adaptive beam formingtechniques are known to combat these problems. But adaptive beam formingfor transmission at 10 GHz or higher frequencies requires massivelyparallel utilization of A/D and D/A converters.

To provide a beamforming system compatible with semiconductor processes,the applicant has provided a number of integrated antenna circuits. Forexample, U.S. application Ser. No. 11/141,283 discloses a beamformingsystem in which an RF signal is distributed through a transmissionnetwork to integrated antenna circuits that include a beamformingcircuit that adjusts the phase and/or the amplitude of distributed RFsignal responsive to control from a controller/phase manager circuit. Ina receive configuration, each beamforming circuit adjusts the phaseand/or the amplitude of a received RF signal from the correspondingintegrated circuit's antenna and provide the resulting adjusted receivedRF signal to the transmission network. Although such integrated antennacircuits consume a relatively small amount of power, transmission lossis incurred through the resulting RF propagation in the transmissionnetwork. To account for such loss, U.S. application Ser. No. 11/141,283discloses a distributed amplification system such that RF signalspropagated through the transmission network are actually amplifiedrather than attenuated. However, the transmission network introducesdispersion as well.

To avoid the dispersion introduced by an RF transmission network, analternative integrated circuit (which may also be denoted as anintegrated oscillator circuit) has been developed such as disclosed inU.S. Pat. No. 6,982,670. For example, each integrated oscillator/antennacircuit may include an oscillator such as a phase-locked loop (PLL) anda corresponding antenna and mixer. In such an embodiment, each PLL isoperable to receive a reference signal and provide a frequency-shiftedsignal output signal that is synchronous with the reference signal.Should an integrated oscillator/antenna circuit be configured fortransmission, its output signal is upconverted in the unit's mixer andthe upconverted signal transmitted by the corresponding antenna.Alternatively, should an integrated oscillator/antenna circuit beconfigured for reception, a received RF signal from the unit's antennais downconverted in the mixer responsive to the frequency-shifted outputsignal from the PLL. Although the integrated oscillator circuit approachdoes not have the dispersion issues resulting from propagation through atransmission network, the inclusion of an oscillator in each integratedoscillator circuit demands significantly more power than thetransmission network approach.

Accordingly, there is a need in the art for beamforming systemscompatible with semiconductor manufacturing processes having reducedpower demands and reduced signal dispersion.

SUMMARY

In accordance with an aspect of the invention, an integrated circuitantenna array includes: a substrate, a plurality of antennas adjacentthe substrate; and an RF network adjacent the substrate, the RE feednetwork coupling to a distributed plurality of amplifiers integratedwith the substrate, wherein the RF feed network and the distributedplurality of amplifiers are configured to form a resonant network suchthat if a timing signal is injected into an input port of the RFnetwork, the resonant network oscillates to provide a globallysynchronized RF signal to each of the antennas.

In accordance with another aspect of the invention, an antenna array isprovided that includes: a semiconductor substrate having a first surfaceand an opposing second surface; a plurality of heavily-doped contactregions extending from the first surface to the second surface; aplurality of antennas formed on an insulating layer adjacent the firstsurface, each antenna being coupled to corresponding ones of the contactregions by vias; and a conductor-based RF feed network adjacent thesecond surface for coupling an input port to the plurality of antennas,the RF feed network coupling to a distributed plurality of amplifiersintegrated into the second surface of the substrate, wherein the RF feednetwork and the distributed plurality of amplifiers are configure toform a resonant network such that if a timing signal is injected intothe input port of the RF feed network, a globally synchronized RF signalis received at each of the antennas.

The invention will be more fully understood upon consideration of thefollowing detailed description, taken together with the accompanyingdrawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a beamforming antenna array in which thebeamforming is performed in the RF domain.

FIG. 2 is a schematic illustration of an RF beamforming interfacecircuit for the array of FIG. 1.

FIG. 3 a is an illustration of a wafer scale resonant transmittingnetwork in accordance with an embodiment of the invention.

FIG. 3 b is an illustration of a wafer scale receiving network havinglinear amplification in accordance with an embodiment of the invention.

FIG. 4 illustrates a wafer scale antenna module including thetransmitting network of FIG. 3 a and the receiving network of FIG. 3 b.

FIG. 5 is a schematic illustration of a matching amplifier in accordancewith an embodiment of the invention.

FIG. 6 is a schematic illustration of a driving amplifier fordistributed amplification in accordance with an embodiment of theinvention.

FIG. 7 illustrates a distributed amplification arrangement with respectto a splitting junction in accordance with an embodiment of theinvention.

FIG. 8 illustrates a distributed amplification arrangement with respectto a splitting junction in accordance with an embodiment of theinvention.

FIG. 9 illustrates a distributed amplification arrangement with respectto a combining junction in accordance with an embodiment of theinvention.

FIG. 10 is a schematic illustration of a matching amplifier for acombining junction used in distributed amplification in accordance withan embodiment of the invention.

FIG. 11 is a cross-sectional view of an integrated antenna circuithaving a coplanar waveguide RF feed network in accordance with anembodiment of the invention.

FIG. 12 is a graph of the resonant oscillation period for a resonanttransmission network with distributed oscillation as a function of thenumber of distributed amplifiers.

FIG. 13 is a schematic illustration of an RF beamforming interfacecircuit adapted to couple to the resonant transmission network of FIG.3.

FIG. 14 is a conceptual illustration of a wafer scale antenna modulehaving a differential distributed oscillator.

Embodiments of the present invention and their advantages are bestunderstood by referring to the detailed description that follows. Itshould be appreciated that like reference numerals are used to identifylike elements illustrated in one or more of the figures.

DETAILED DESCRIPTION

Reference will now be made in detail to one or more embodiments of theinvention. While the invention will be described with respect to theseembodiments, it should be understood that the invention is not limitedto any particular embodiment. On the contrary, the invention includesalternatives, modifications, and equivalents as may come within thespirit and scope of the appended claims. Furthermore, in the followingdescription, numerous specific details are set forth to provide athorough understanding of the invention. The invention may be practicedwithout some or all of these specific details. In other instances,well-known structures and principles of operation have not beendescribed in detail to avoid obscuring the invention.

The present invention provides a wafer scale antenna module in which aresonant transmission network with distributed amplification is drivenby a triggering pulse waveform such that the entire transmission networkoscillates acting as a distributed oscillator. Advantageously, the RFsignal from the resulting distributed oscillator thereby arrivessynchronously at a plurality of integrated antenna circuits coupled tothe distributed oscillator. Each integrated antenna circuit may includea modulator such as the pulse shaping circuit disclosed in U.S.application Ser. No. 11/454,915. In this fashion, ultra wideband pulsesmay be propagated from the resulting wafer scale antenna module withoutincurring dispersion caused by propagation of the pulses through atransmission network. Significantly, however, such ultra wideband pulsesmay be generated without the need for oscillators such as a voltagecontrolled oscillator (VCO) in each integrated antenna circuit. Thus,the disclosed distributed oscillator provides substantial power savingsas opposed to integrated oscillator circuit embodiments.

Embodiments of the wafer scale beamforming approach disclosed herein maybe better understood with reference to the beamforming system of FIG. 1,which illustrates an integrated RF beamforming and controller unit 130.In this embodiment, the receive and transmit antenna arrays are the samesuch that each antenna 170 functions to both transmit and receive. Aplurality of integrated antenna circuits 125 each includes an RFbeamforming interface circuit 160 and receive/transmit antenna 170. RFbeamforming interface circuit 160 adjusts the phase and/or the amplitudeof the received and transmitted RF signal responsive to control from acontroller/phase manager circuit 190. Although illustrated having aone-to-one relationship between beamforming interface circuits 160 andantennas 170, it will be appreciated, however, that an integratedantenna circuit 125 may include a plurality of antennas all driven by RFbeamforming interface circuit 160.

A circuit diagram for an exemplary embodiment of RF beamforminginterface circuit 160 is shown in FIG. 2. Note that the beamformingperformed by beamforming circuits 160 may be performed using eitherphase shifting, amplitude variation, or a combination of both phaseshifting and amplitude variation. Accordingly, RF beamforming interfacecircuit 160 is shown including both a variable phase shifter 200 and avariable attenuator 205. It will be appreciated, however, that theinclusion of either phase shifter 200 or attenuator 205 will depend uponthe type of beamforming being performed. To provide a compact design, RFbeamforming circuit may include RF switches/multiplexers 210, 215, 220,and 225 so that phase shifter 200 and attenuator 205 may be used ineither a receive configuration or a transmit configuration. For example,in a receive configuration RF switch 215 routes the received RF signalto a low noise amplifier 221. The resulting amplified signal is thenrouted by switch 220 to phase shifter 200 and/or attenuator 205. Thephase shifting and/or attenuation provided by phase shifter 200 andattenuator 205 are under the control of controller/phase manager circuit190. The phase shifted signal routes through RF switch 225 to RF switch210. RF switch 210 then routes the signal to IF processing circuitry(not illustrated).

In a transmit configuration, the RF signal received from IF processingcircuitry (alternatively, a direct down-conversion architecture may beused to provide the RF signal) routes through RF switch 210 to RF switch220, which in turn routes the RF signal to phase shifter 200 and/orattenuator 205. The resulting shifted signal is then routed through RFswitch 225 to a power amplifier 230. The amplified RF signal then routesthrough RF switch 215 to antenna 170 (FIG. 1). It will be appreciated,however, that different configurations of switches may be implemented toprovide this use of a single set of phase-shifter 200 and/or attenuator205 in both the receive and transmit configuration. In addition,alternate embodiments of RF beamforming interface circuit 160 may beconstructed not including switches 210, 220, and 225 such that thereceive and transmit paths do not share phase shifter 200 and/orattenuator 205. In such embodiments, RF beamforming interface circuit160 would include separate phase-shifters and/or attenuators for thereceive path and transmit path.

To assist the beamforming capability, a power detector 250 functions asa received signal strength indicator to measure the power in thereceived RF signal. For example, power detector 250 may comprise acalibrated envelope detector. As seen in FIG. 1, a power manager 150 maydetect the peak power determined by the various power detectors 250within each integrated antenna circuit 125. The integrated antennacircuit 125 having the peak detected power may be denoted as the“master” integrated antenna circuit. Power manager 150 may thendetermine the relative delays for the envelopes for the RF signals fromthe remaining integrated antenna circuits 125 with respect to theenvelope for the master integrated antenna circuit 125. To transmit inthe same direction as this received RF signal, controller/phase manager190 may determine the phases corresponding to these detected delays andcommand the transmitted phase shifts/attenuations accordingly.Alternatively, a desired receive or transmit beamforming direction maysimply be commanded by controller/phase manager 190 rather than derivedfrom a received signal. In such embodiment, power managers 150 and 250need not be included since phasing information will not be derived froma received RF signal.

Regardless of whether integrated antenna circuits 125 perform theirbeamforming using phase shifting and/or amplitude variation, theshifting and/or variation is performed on the RF signal received eitherfrom the IF stage (in a transmit mode) or from its antenna 170 (in areceive mode). By performing the beamforming directly in the RF domainas discussed with respect to FIGS. 1 and 2, substantial savings areintroduced over a system that performs its beamforming in the IF orbaseband domain. Such IF or baseband systems must include A/D convertersfor each RF channel being processed. In contrast, the system shown inFIG. 1 may supply a combined RF signal from an adder 140. From an IFstandpoint, it is just processing a single RF channel for the system ofFIG. 1, thereby requiring just a single A/D. Accordingly, the followingdiscussion will assume that the beamforming is performed in the RFdomain. The injection of phase and/or attenuation control signals bycontroller/phase manager circuit 190 into each integrated antennacircuit 125 may be performed inductively as discussed incommonly-assigned U.S. Pat. No. 6,885,344, the contents of which areincorporated by reference.

A transmission network couples the RF signal from the IF stage (oralternatively, from a baseband stage in a direct downconversionembodiment) to the RF beamforming interface circuits. As set forth inU.S. application Ser. No. 11/141,283, a particularly advantageoustransmission network with regard to a wafer scale approach is a coplanarwaveguide (CPW) network. Although the scope of the invention includesthe use of any suitable architecture for a transmission network such asCPW, microstrip, and planar waveguide, CPW enjoys superior shieldingproperties over microstrip. Thus, the following discussion will assumewithout loss of generality that the transmission network is implementedusing CPW. This network may be arranged in an “H” array such that theelectrical length from an RF input port to any given integrated antennacircuit is the same as that to all the remaining integrated antennacircuits. Although CPW has superior shielding properties, the RFpropagation across a CPW network on a semiconductor wafer such as an 8″wafer may introduce losses as high as 120 dB. To counteract such losses,a plurality of distributed amplifiers may be coupled to the CPW networkas disclosed in U.S. application Ser. No. 11/141,283. For example, afirst linear transistor amplifier (which may be denoted as a drivingamplifier) amplifies a received RF signal into a length of the CPWnetwork into a second linear transistor amplifier (which may be denotedas a matching amplifier) configured to match its output impedance to thecharacteristic impedance of the CPW network. Both the gain of thedriving amplifier and the gain and the output impedance of the matchingamplifier are tuned using reactive loads such as integrated inductors.In this fashion, resistive losses are minimized. These gains aremaintained so that linear operation is achieved. In this fashion, an RFsignal driven into an input port of the CPW network is linearlyamplified and propagated to the integrated antenna circuits, despite thetransmission line losses.

In the present invention, it has been observed that the combination ofthe resulting active devices and the transmission network can be tunedto form a resonant network. Because the network is resonant, aglobally-synchronized oscillation can be induced by triggering thenetwork with an appropriate timing signal. The distributed amplifiersthus injection lock to each other such that the resonant network forms adistributed oscillator providing each integrated antenna circuit with aglobally synchronized RF signal. This RF signal may then be modulated ifdesired such as through the pulse shaping circuit of U.S. applicationSer. No. 11/454,915 (the contents of which are incorporated byreference) or through an alternative pattern generation. In addition,the RF signal received at the integrated antenna circuits may be phaseshifted using a phase shifter such as the analog phase shifter in U.S.application Ser. No. 11/535,928 (the contents of which are incorporatedby reference) or any other suitable phase shifter such as disclosed inU.S. application Ser. No. 11/182,344.

Turning now to FIG. 3 a, a resonant half-duplex transmission network 410is implemented in an 8″ wafer scale antenna module 400 having 64 antennaelements 170. The triggering signal to trigger the resonant oscillationis injected into a center feed point 405. Distributed amplifiers 430coupled to the network then injection lock to each other such that eachantenna 170 may receive a globally synchronized RF signal. In contrastto the transmission network, a half-duplex receiving CPW network 510 asseen in FIG. 3 b for wafer scale antenna module 400 operates in thelinear amplification regime as described earlier. A wafer scale antennamodule 500 including 256 antenna elements 170 is shown in FIG. 4 thatincludes both a resonant transmitting network 400 and a linearamplification receiving network 510 is illustrated in FIG. 5. Because ofthe global synchronization provided by the resonant operation of network400, it may also be denoted as a central clock distribution network.

Each transmission network may be single-ended or differential. In oneembodiment, the network may comprise a coplanar waveguide (CPW) having aconductor width of a few microns (e.g., 4 microns). With such a smallwidth or pitch to the network, a first array of 64 antenna elements anda second array of 1024 antenna elements may be readily networked in an 8inch wafer substrate for 10 GHz and 40 GHz operation, respectively.Alternatively, a wafer scale antenna module may be dedicated to a singlefrequency band of operation.

The design of the distributed amplifiers is not critical so long as theyprovide sufficient amplification and achieve a resonant operation withthe transmission network. Thus, it will be appreciated that thedistributed amplifiers may comprise the driving/matching amplifiersdescribed earlier or alternative distributed amplifiers may be used. Inone embodiment, a driving amplifier in the receiving and transmissionnetworks is followed by a matching amplifier for efficient performance.An exemplary embodiment of a FET-based matching amplifier 600 isillustrated in FIG. 5. Matching amplifier 600 couples to a coplanarwaveguide network (not illustrated) at input port Vin and output portVout. An analogous BJT-based architecture may also be implemented. TheFETs may be either NMOS or PMOS. A first NMOS FET Q1 605 has its draincoupled through an integrated inductor (L1) 610 to a supply voltage Vcc.This integrated inductor L1 may be formed using metal layers in asemiconductor process as discussed in commonly-assigned U.S. Pat. No.6,963,307. Because such an integrated inductor L1 will also have a straycapacitance and resistance, these stray effects are modeled by capacitorC1 and resistor R1. The metal layers in the semiconductor process mayalso be used to form a DC blocking capacitor CS and an output capacitorC_(out). The supply voltage also biases the gate of Q1. Q1 has its draindriving Vout and its drain coupled to a second NMOS FET Q2 620. Avoltage source 630 coupled through a high value resistor or configuredtransistor biases the gate of Q2 620 with a voltage Vgb (whereas in aBJT embodiment, the base of Q1 is biased by a current source). Thesource of Q2 620 couples to ground through an integrated inductor (L2)640. Analogous to inductor 610, inductor 640 has its stray capacitanceand resistance modeled by capacitor C2 and resistor R2. It may be shownthat an input resistance Rin for amplifier 600 is as follows:Rin=(gm)*L2/Cgswhere gm is the transconductance for Q2 620, L2 is the inductance of theinductor 640 and Cgs is the gate-source capacitance for Q2 620. Thus, Q2620 and inductor 640 characterize the input impedance and may be readilydesigned to present a desired impedance. For example, if an inputresistance of 50Ω is desired (to match a corresponding impedance of theCPW network), the channel dimensions for Q2 and dimensions for inductor640 may be designed accordingly. The gain of matching amplifier 600 isproportional to the inductance of L1.

An exemplary driving amplifier 700 is illustrated in FIG. 6. Drivingamplifier 700 is constructed analogously to matching amplifier 600except that no inductor loads the source of Q2 705 (alternatively, aninductor having a fraction to 1/10 the inductance of L1 may load thesource of Q2). The gain of driving amplifier 700 is proportional to theinductance of L1. A transistor Q1 710 has its drain loaded withintegrated inductor L1 715 in a similar fashion as discussed with regardto Q1 605 of matching amplifier 600. Inductor 715 determines a centerfrequency Fd for driving amplifier 700 whereas both inductors 640 and610 establish a resonant frequency Fm for matching amplifier 600. It maybe shown that the band-pass center frequency Fc of a series-connecteddriving and matching amplifier is given asFc=½*sqrt(Fd ² +Fm ²)

Referring back to FIG. 3 a, a series of driving amplifier/matchingamplifier pairs 430 are shown coupling feed point 405 to a first networkintersection 460. In such an “H” configured network array, network 410will continue to branch from intersection 460 such as at an intersection470. For a half-duplex embodiment, driving amplifier/matching amplifierpairs 430 may also be incorporated in receiving network 510 as seen inFIGS. 3 b and 4. For illustration clarity, the distribution of thedriving amplifier/matching amplifier pairs 430 is shown only in selectedtransmission paths in FIGS. 3 a and 3 b. It will be appreciated thatboth the driving amplifiers and the matching amplifiers may beconstructed using alternative arrangements of bipolar transistors suchas PNP bipolar transistors or NPN bipolar transistors. In a bipolarembodiment, biasing voltage sources 630 are replaced by biasing currentsources. In addition, the RF feed network and these amplifiers may beconstructed in either a single ended or differential fashion. DC andcontrol lines may be arranged orthogonally to the RF distributiondirection for isolation. In addition, this same orthogonality may bemaintained for the RF transmit and receive networks in a full duplexdesign.

Turning now to FIG. 7, a single driving amplifier/matching amplifierpair 430 may both precede and follow network branching intersections800, 805, and 810 in transmission network 410. Alternatively, just asingle pair 430 may drive each branching intersection. It will beappreciated that the same considerations apply to a receiving (and hencecombining) network.

The resonant network properties are influenced by the distance betweendriving amplifiers and matching amplifiers in successive drivingamplifier/matching amplifier pairs. For example, as seen for RF networkportion 900 in FIG. 8, its input or source is received at a first driveramplifier 700 a, which drives a matching amplifier 600 a separated fromdriver 700 a by a length of network transmission line (such as coplanarwaveguide) of length TL1. Driver amplifier 700 a and matching amplifier600 a thus constitute a first driving amplifier/matching amplifier pair530 a, which may also be denoted as a load balanced amplifier (LBA).Matching amplifier 600 a is immediately followed by a driver amplifier700 b, which couples to the output of matching amplifier 600 a directlyin the active circuitry silicon rather than through a transmission linesection. In this fashion, die space on the wafer substrate is conserved.However, it will be appreciated that an RF network CPW transmission linesegment could also be used to couple matching amplifier 600 a to drivingamplifier 700 b. Driver amplifier 700 b drives a matching amplifier 600b separated from driver 700 b by a length TL2 of network transmissionline. Driver amplifier 700 b and matching amplifier 600 b thus form asecond driving amplifier/matching amplifier 530 b. The necessary biasingand inductance loading as described with respect to FIGS. 5 and 6 arerepresented by bias and filter impedances 910. In general, the sum ofTL1 and TL2 should equal one half of the center frequency wavelength. Bychanging the ratio of TL1/TL2 and the output capacitance, a maximumstable gain of approximately 20 to 30 dB may be obtained for 10 GHz to,for example, 40 GHz operation. In a linear amplification (as opposed toresonant operation) 10 GHz embodiment, stable gain and frequencyperformance may be realized for a capacitance load of 50 fF as TL1/TL2is varied from 40% to 80%.

In prior art RF distribution networks splitting and combining signalswas problematic and involved cumbersome combiner or splitter circuitry.However, note the simplicity involved for the coupling of matchingamplifier 600 b through a splitting junction 950 to driver amplifiers700 c and 700 d. This coupling occurs through a node in the activecircuitry substrate to conserve wafer substrate area. However, thissubstrate coupling may be replaced by a CPW transmission line segment inalternative embodiments. As compared to prior art splitters, not only isthere no loss coupling through splitting junction 950, but there is again instead. Moreover, transmission through the RF feed network is lowloss and low noise because the driver and matching amplifiers are tunedwith reactive components only—no resistive tuning (and hence loss) needbe implemented.

The same low loss and simplicity of design advantages are present withrespect to combining junction 1000, 1005, and 1010 for a receivingnetwork as seen in FIG. 9. For example, with respect to junction 1000,two combiner matching amplifiers 1020 and 1025 (discussed further withregard to FIG. 10) couple through a node in the active circuitrysubstrate to a driving amplifier 700 e to conserve wafer substrate area.However, it will be appreciated that a CPW transmission line segment maybe used to perform this coupling in alternative embodiments. Bias andfilter impedance 910 is thus shared by both combiner matchingamplifiers.

Turning now to FIG. 10, a combiner matching amplifier 1101 isdistinguished from a non-combiner matching amplifier such as discussedwith respect to FIG. 5 by the absence of L1 at the drain of a FET Q11100. A FET Q2 1105 has its drain loaded by the matching inductor 640for impedance matching as discussed with respect to FIG. 5. A commonload inductor (not illustrated) couples to output node Vout to uniformlyload all the involved combiner matching amplifiers.

The integration of the CPW network and the distributed amplificationinto a wafer scale integrated antenna module (WSAM) may be betterunderstood by classifying the WSAM into three layers. The first layerwould be a semiconductor substrate, such as silicon. On a first surfaceof the substrate, antennas such as patches for the integrated antennacircuits are formed as discussed, for example, in U.S. Pat. No.6,870,503, the contents of which are incorporated by reference herein.Active circuitry for the corresponding integrated antenna circuits thatdrive these antennas are formed on a second opposing surface of thesubstrate. The CPW transmission network is formed adjacent this secondopposing surface. The second layer would include the antennas on thefirst side of the substrate whereas the third layer would include theCPW network. Thus, such a WSAM includes the “back side” featuredisclosed in U.S. Ser. No. 10/942,383, the contents of which areincorporated by reference, in that the active circuitry and the antennasare separated on either side of the substrate. In this fashion,electrical isolation between the active circuitry and the antennaelements is enhanced. Moreover, the ability to couple signals to andfrom the active circuitry is also enhanced. As discussed in U.S. Ser.No. 10/942,383, a heavily doped deep conductive junction through thesubstrate couples the active circuitry to vias/rods at the firstsubstrate surface that in turn couple to the antenna elements. Formationof the junctions is similar to a deep diffusion junction process usedfor the manufacturing of double diffused CMOS (DMOS) or high voltagedevices. It provides a region of low resistive signal path to minimizeinsertion loss to the antenna elements.

Upon formation of the junctions in the substrate, the active circuitrymay be formed using standard semiconductor processes. The activecircuitry may then be passivated by applying a low temperature depositedporous SiOx and a thin layer of nitridized oxide (Si_(x)O_(y)N_(z)) as afinal layer of passivation. The thickness of these sealing layers mayrange from a fraction of a micron to a few microns. The opposing secondsurface may then be coated with a thermally conductive material andtaped to a plastic adhesive holder to flip the substrate to expose thefirst surface. The substrate may then be back ground to reduce itsthickness to a few hundreds of micro-meters.

An electric shield may then be sputtered or alternatively coated usingconductive paints on background surface. A shield layer over theelectric field may form a reflective plane for directivity and alsoshields the antenna elements. In addition, parts of the shield formohmic contacts to the junctions. For example, metallic lumps may bedeposited on the junctions. These lumps ease penetration of the via/rodsto form ohmic contacts with the active circuitry.

In an alternative embodiment, the CPW network may be integrated on theantenna side of the substrate. Because the backside approach has theisolation and coupling advantages described previously, the followingdiscussion will assume without loss of generality that the RF feednetwork is integrated with the substrate in a backside embodiment. Forexample as seen in cross-section in FIG. 11, a semiconductor substrate1201 has opposing surfaces 1202 and 1203. Antenna elements 1205 such aspatches are formed on a dielectric layer 1206 adjacent to surface 1202.Active circuitry 1210 integrated with substrate 301 includes the drivingand matching amplifiers for an RF feed network 1204 having CPWconductors S1 and S2. Adjacent surface 303, metal layer M1 includesinter-chip and other signal lines. Metal layer M2 forms, among otherthings, a ground plane for CPW conductors S1 and S2, which are formed inmetal layer 5 as well as ground plates 1220. Metal layer M4 provides aconnecting layer to couple CPW conductors together as necessary. Thedriving and matching amplifiers within active circuitry 1210 couplethrough vias (not illustrated) in apertures in the ground plane in metallayer M2 to CPW conductors S1 and S2. This active circuitry may alsodrive antennas 1205 through a plurality of vias 1230 that extend throughthe dielectric layer. An electric shield layer 1240 isolates thedielectric layer from surface 1202 of the substrate. The antennas may beprotected from the elements and matched to free space through apassivation layer.

Just as active circuitry is distributed across the CPW network foramplification (using, e.g., the matching and driving amplifiersdiscussed previously), active circuitry may also be used to formdistributed phase shifters as will be explained further herein. Thelocation of the distributed phase shifters depends upon the granularitydesired for the beam steering capability. For example, referring back toFIGS. 3 a and 3 b, each antenna element 170 could receive individualphase shifting through an adjacent and corresponding distributed phaseshifter. To save costs and reduce power consumption, subsets of antennaelements 170 may share in the phase shifting provided by a correspondingdistributed phase shifter. For example, consider a subset 450 or 550having sixteen antenna elements 170. As seen in FIG. 3 a, a distributedphase shifter located adjacent an intersection 460 of network 410 wouldprovide equal phase shifting for each of the elements within a subset450. Similar subsets would have their own distributed phase shifter.Similarly, as seen in FIG. 3 b, a distributed phase shifter locatedadjacent an intersection 560 of network 510 would provide equal phaseshifting for each of the elements within subset 550 with respected areceived RF signal. Thus, it may be appreciated that the granularity ofthe beam steering capability is a design choice and depends upon desiredmanufacturing costs and associated complexity.

It is believed that the resonant frequency of the resonant transmissionnetwork depends on the number of the distributed amplifiers (entirelength of transmission line) from central triggering point 405 reachingto each individual integrated antenna circuit. For example, it isbelieved that resonant oscillation may be achieved for a 128 quarterwavelength transmission distance from point 405 to each integratedantenna circuit with TL1=400 micron and TL2=1250 micron, a Q1 currentsink ability of 15× that of Q2 (in both driver and matching amplifier)with 2× source ability and a triggering pulse width of 20 pS andrepetition rate of 3600 pS produces a steady state oscillation of 600 mVand frequency of 20 GHz at the termination point for appropriate valuesof the resonant loads. Advantageously, such 20 GHz distribution may needconsume only 30 mV across the wafer. In contrast, an integratedoscillator circuit approach may require 1000 times more power. Changingthe pulse triggering repetition to 400 pS and reducing the load to 3×and the sink to 3× with regard to the minimum geometry for the Q1 and Q2transistors yields a 33 GHz oscillation frequency. Further reduction oftransistor Q1 to 1× and Q2 to 1× results in a frequency of oscillationclose to 45 GHz. In general, as the number of distributed amplifiers inincreased in the resonant network, the resonant oscillation period willincrease due to the parasitic loading from the increased number ofactive devices as shown in FIG. 12.

An exemplary RF beamforming interface circuit 160 configured to coupleto the resonant transmission network (distributed oscillator) isillustrated in FIG. 13. Because of the global synchronization (GS)provided by the distributed oscillator (DO), the distributed oscillatormay also be denoted as a distributed oscillator for globalsynchronization (DOGS) 1090. The RF signal from DOGS 1090 couples to apattern generator 1080 such as the pulse shaper discussed earlier.Alternatively, any suitable modulation may be imposed on the RF signalin pattern generator 1080 as controlled by control unit 190. Theremaining components in unit 160 function as discussed with regard toFIG. 2 to provide a received signal to a receiving network 1050.

As discussed earlier, the transmission network may be constructed aseither a single ended or a differential network. A differential DOGS1425 is illustrated in FIG. 14. It can be shown that a differential DOGS1425 provides a significant signal-to-noise advantage over a singleended design. DOGS 1425 is triggered by a differential triggering signal1415 which results in a globally synchronized oscillation such that anRF signal is delivered to RF beamforming interface circuits 160.

It will be obvious to those skilled in the art that various changes andmodifications may be made without departing from this invention in itsbroader aspects. The appended claims encompass all such changes andmodifications as fall within the true spirit and scope of thisinvention.

1. An integrated circuit antenna array, comprising: a substrate, aplurality of antennas adjacent the substrate; and an RF network adjacentthe substrate, the RF feed network coupling to a distributed pluralityof amplifiers integrated with the substrate, wherein the RF feed networkand the distributed plurality of amplifiers are configured to form aresonant network such that if a timing signal is injected into an inputport of the RF network, the resonant network oscillates to provide aglobally synchronized RF signal to each of the antennas.
 2. Theintegrated circuit antenna array of claim 1, wherein the substrate is asemiconductor wafer substrate.
 3. The integrated circuit antenna arrayof claim 1, wherein the RF feed network is implemented using waveguidesselected from the group consisting of microstrip waveguides, co-planarwaveguides, and planar waveguides.
 4. The integrated circuit antennaarray of claim 3, wherein the antennas are adjacent a first surface ofthe substrate and wherein the RF feed network is a co-planar waveguidenetwork adjacent an opposing surface of the substrate, the activecircuitry being integrated into the opposing surface.
 5. The integratedcircuit antenna array of claim 3, wherein the co-planar waveguidenetwork is formed in metal layers adjacent the opposing surface of thesubstrate.
 6. An integrated RF feed network, comprising: a substrate; awaveguide network formed adjacent to a surface of the substrate, whereinthe substrate includes a plurality of driving amplifiers and matchingamplifiers arranged in pairs, the waveguide network being segmented intotransmission line segment pairs, each transmission line segment pairhaving a first segment of length TL1 and a second segment of length TL2,each segment corresponding to a driving amplifier and matching amplifierpair, wherein for each segment the corresponding driving amplifieramplifies the RF signal through the segment to the correspondingmatching amplifier, and wherein the waveguide network and the amplifierpairs are configured to form a resonant network such if a timing signalis injected into an input port of the waveguide network, the resonantnetwork oscillates to provide a globally synchronized RF signal at aplurality of output ports.
 7. The integrated RF feed network of claim 6,wherein each driving amplifier includes an inductor having an inductanceL1D, and wherein each matching amplifier has a first inductor having aninductance L1M and a second inductor having an inductance L2M, eachmatching amplifier being configured such that its input impedancedepends upon inductance L3.
 8. The integrated RF feed network of claim7, wherein for each transmission line segment pair, the values ofinductances L1D, L1M, L2M and lengths TL1 and TL2 are chosen to achievea resonant operation.
 9. The integrated RF feed network of claim 8,wherein each inductor is formed in metal layers adjacent to the surfaceof the substrate.
 10. The integrated RF feed network of claim 9, whereinthe RF feed network is a coplanar waveguide network formed in the metallayers.
 11. An antenna array, comprising: a semiconductor substratehaving a first surface and an opposing second surface; a plurality ofheavily-doped contact regions extending from the first surface to thesecond surface; a plurality of antennas formed on an insulating layeradjacent the first surface, each antenna being coupled to correspondingones of the contact regions by vias; and a conductor-based RF feednetwork adjacent the second surface for coupling an input port to theplurality of antennas, the RF feed network coupling to a distributedplurality of amplifiers integrated into the second surface of thesubstrate, wherein the RF feed network and the distributed plurality ofamplifiers are configure to form a resonant network such that if atiming signal is injected into the input port of the RF feed network, aglobally synchronized RF signal is received at each of the antennas. 12.The antenna array of claim 11, wherein the RF feed network is segmentedinto transmission line segment pairs, each transmission line segmentpair having a first segment of length TL1 and a second segment of lengthTL2, the distributed amplifiers being organized into driving amplifiersand matching amplifiers, each segment corresponding to a drivingamplifier and matching amplifier pair, wherein for each segment thecorresponding driving amplifier drives RF signal through the segment tothe corresponding matching amplifier.
 13. The antenna array of claim 12,wherein the lengths TL1 and TL2 are chosen to maximize gain at a desiredoperating frequency.
 14. The antenna array of claim 15, wherein the RFfeed network comprises a coplanar waveguide network.